Band pass filter



April 7, 1942.

N. M.' RusT ErAL BAND PASS FiLTER Filed oct. 1, 1940 8 Sheets-SheetI 1 ai .DA'rfc To@ I'NVENTORS N. M. RUST J. D- BRA/LGFORD A15ril7,v1942. A N M RUST TAL 2,278,801

` BAND PAss FILTER Filed Oct. l, 1940 8 Sheets-'Sheet 2 AAAAAAA vvvvvvvv ATTQRNEY J-D. BRAILSFORD April 7, 1942. N. M. RUST ETAL Y BAND PAss FILTER Filed Oct. l, 1940 0 +10 AMM 8 Sheets-Sheet l5 NVENTORSl N.M. RUST J- 0- BRLSFRD E. F. GOQOUGH 7 t ATTORNEY April 7, 1942. N. M. RUST ETAL 2,278,801'

' BAND PAss FILTER Filedpct. 1, 1940 v a sheets-sheet 4 ffyj; [Finn/FMR of? 5 Tic TOR AAAAAL vvv AAAAAAAAIAAAA.. Vvvvvvvvvvvvvv April 7, 1942. N. M. RUsTE-r AL 2,278,801

BAND PAss NLTER v Filed Oct. l, 1940 8 Sheets-Sheet 5 I N VEN TORS N. M. RUST J. D- BRA/LSFORD. E. F. GOODENOUGH TT'ORNEY April 7, 1942.. N. M. RUST ET Al. A 2,278,801

BAND PASS FILTER Filed Oct. l, 1940 8 Sheets-Sheet 6 V UWENTORS N. M. @usr J. D. BRA ILS/:ORD

E. F. 60ans/vous# l April 7171942. N. M. RUST ET Al. 2,278,801

BAND PASIS FILTER Filed oct. 1, 1940 s sheets-sheet '7 T'TORNEY April 7, 1942. N; M. RUST ETAL 2,278,801

BAND PASS FILTER Filed4 oot. 1, 1940 v23 Sheets-Sheet 8 IN VEN T S N. M.' RUS J. D. BRAILSFORD E. F. GOODENUUG/ A TTORNEY Patented Apr. 7, 1942 n UNITED `STATES VAT'ENT OFFICE l Nol Meyer Rust,

Brailsford, London,

land,

Chelmsford, Joseph Douglas and Ernest Frederick Goodenough, Springfield, Chelmsford, Engassignors to Radio Corporation of America, a. corporation of Delaware Application October 1, 1940, Serial No. 359,182 In Great Britain June 8, 1939 13 Claims.

This invention relates to band pass filters and has for its object to'provide improved filters which are comparatively iiexible in design and will facilitate the production of (1) wide band amplifiers of high gain and large signal-to-noise ratio (2) narrow band amplifiers of greatselec-V tivity and with a fiat topped response curve such asfwill give a good signal-to-noise ratio or (3*) variable selectivity amplifiers in which selectivity and fidelity can be readily adjusted inaccordance with the operating conditions at theL time. Though not limited to its application thereto the primary applications of the invention are to television amplifiers and to intermediate frequency amplifiers for broadcast and like receivers.

According to this invention a band pass filter effect is obtained `by superimposing upon the frequency-impedance characteristics of one network, the frequency-impedance characteristic .of at least one further network which is coupled to the first in such manner that the impedance elements in the second network are, in effect, re-

versed. In other words, there are employed, in

accordance with this invention, two networks coupled together in such manner that the second .network acts as though it were directly included in and were part of the first network but with its impedance elements reversed. The .expression reversed as employed in this specification and in the claims, is used with a somewhat special meaning which will be explained hereinafter.

The invention is illustrated in and explained in connection with the drawings which accompany the present specification, wherein Fig. 1 is a circuit in diagrammatic form which serves to explain certain aspects of the invention; Fig. 2 shows a wide band high gain amplier circuit in accordance with the invention; Fig. 3 shows the equivalent grid circuit impedance of the first valve of Fig. 2; Fig. 4 is alsecond equivalent circuit under certain conditions; Figs. 5 and 6 are frequency-impedance characteristics obtained from the circuit of Fig. 2; Figs. 78, 9 and 13 are circuit elements which serve to explain the invention as applied to narrow band, highly selective band-pass filters; Figs. 10, ll'and 12 arev plain the utilization of stray capacities present in television carrier or intermediate frequency `cir.- cuits; Figs. 20 and 21k serve to explain certain methods of overcoming phase shift between` the output and input voltages yof a high frequencyv amplifier; Figs. -22 to y26 show practical circuit arrangements for Ycompensating for-phase shift; Figs. 2'?, 28 and 29 `sl'iownetworks that may constitute the coupling impedance Z1 of Fig. 20;:Fig.

30 is a circuit equivalent to that of Fig. 29.; and

Figs. 31 and 32 are circuits according to the invention disclosing certain modifications'.

In order that the invention `may be the better understood there will first begiven .a brief and simplified, description of the phenomenacupo which the invention is based.

Consider first a :circuit as shown in Figure 1, which is an explanatory diagrammatic figure. In this circuit there are: two valves Vl, V2, the'plate of the first of whichy is coupled to the grid of the second by an impedance of A-valve Zn which is equal to n A where Yn is the corresponding admittance. The

eifective admittanceappearing across the terminals AB of Figure 1, due to the .action of the circuit, is taken as Y1 which is equal to ,l 'Z where Z1 is the corresponding impedance. 'Ihe 4anode of the second valve is capacity coupled to the grid ofthe first. The extrenal plate impedance Z of the second valve is lvery'large as compared to Z1.

Neglecting effects due to stray couplings and n the like,l then Ylz-yi'gz `ZU 1 ZI- glgzy where g1 and g2 are the mutual conductances of the first and second valves respectively.r The relationships above set forth maybe summarised as a Vtheorem thus: the admittance thrown back to the terminals AB (see Figure '1) follows the same law of variation with frequency as the impedance at the gridy of the ,second valve but is reversed in sign. vThe'impedance thrown across A, B, although f negative, does not` 'follow vthe same law as the impedance .Znbutffollow's the same law vas its reciprocal Yu; '.In otherwords the impedance Zn is in` effectinverted and also reversed in sign. The term reversedfas employed in this specification yis meant to indicate that the impedance acted upon inithis-:manner is at all frequencies proportionate to that 'of its potentially inverse arm (seep. 131"Trar`1smis sion Networks and Filters"by'R.fE. Shea, 1929) the grid of VI.

but is also of opposite sign. The term is specially applied to meanA both inversion and reversal of sign. With such reversal action therefore, the network acted upon is, in effect, equivalent to one which has been derived by the well known Yrules for determining potentially inverse networks the impedance of which however is of opl posit@ sign. Thus a number of elements, or groups of elements, which are in series or parallel in the grid circuit of V2 would appear as a corresponding number in parallel or series in the grid circuit of VI. Moreover, wherever in the grid circuit of V2 an inductance or capacity is used, it may be considered as acting as a negative capacity or inductance inthe grid circuit 'of VI.

Hence the action of any compound circuit may be derived by regarding each element or group of elements in therabove manner, i. e. parallel elements or groups of elements appear as series (and `vice versa), inductance as capacity (and vice versa) and resistance as resistance, the sign .ofthe impedance of the equivalent network being reversed.

The present invention utilises the above described phenomena to enable the characteristics of a tuned circuit or other network to be improved in desired manner by superimposing there- .on characteristics provided by a reversed second network.

Figure 2 shows a wide band high gain amplifier in accordance with this invention. This am- `pliiler comprises two valves VI, V2, a parallel tuned circuit in the grid circuit of the first valve and a network consisting of resistance I, 2, 3, a capacity 4, and an inductance 5, connected as shown in the grid circuit of the second valve. The grid circuit impedance of the first valve is marked Z1 and that of the second valve Zn. The anode of valve V2 is capacity coupled backto work Zn is in effect reversed and the reversed equivalent included in the net Work Z1 so that the equivalent grid circuit impedance of the rst valve may be represented as shown in Figure 3. In Figure-3 the references I, 2, 3, 4, and 5 correspond to the similarly numbered references in Figure 2, and the letters N, superimposed upon With this arrangement the netthe various circuit elements in Figure 3, indicate that those elements are negative. Thus 4N is the negative inductance obtained vby reversing capacity 4 of Figure 2. If the resistance element 2 of Figure 2 be made much greater than the resistance element 3, then in the reversed iin-fl pedance network of Figure 3, 2N will be very much smaller than the impedance of the parallel combination 3N, 4N, 5N, with which it is in series and may be neglected so that the approximate circuit equivalent is as shown in Figure 4. It will be directly seen from Figure 4 that it is possible, by correct choice of circuit constants and suitable adjustment of the operating parameters of the valves VI and V2 (Figure 2) to make the reversed element 4N of Figure 4 neu-v tralise the positive inductance in that figure, the reversed element 5N neutralise the positive capacity and to make the negative resistance 3N compensate for the positive resistance or, applying these remarks to Figure 2, the element 4 in Zn can by reversal be made to neutralise the inductance in Z1, the element 5 in Zn by reversal to neutralise the capacity in Z1, and the resistance 3 in Zn to compensate to an extent determined by considerations of stability for the resistance in Zz. As the eiect of a negative resistance in parallel with a positive resistance is to raise the impedance, the result is achieved that over a range of frequencies dependent upon the relation of the elements 2 and 3 in Figure 2 a substantially constant over-all impedance is obtained. By incorporating such an arrangement in an amplier circuit a level response curve for this range of frequencies can be obtained. At a frequency outside the constant impedance range, where the impedance of the circuit 3, 4, 5, (Figure 2) becomes comparable with that of resistance 2 (Figure 2) the reversed element 2N begins to take effect, the compensating action is destroyed, the net or overall impedance changes and ultimately falls to a low value. Variation of the value of resistance I varies both the overall impedance and the band width, i. e. the range of frequencies over which the impedance remains substantially constant.

The results achieved in an experimentally tested case are represented graphically in the accompanying Figures 5 and 6. In Figure 5 impedance is plotted along lthe ordinate line against frequency in cycles per second. The mid-band frequency is marked O, AF representing changes in frequency away from the mid-band frequency. The curves are drawn for diierent values of the product gl and y2 in mhos, these values being marked on the respective curves. gI and g2 are equal to one another and each is equal to 1 milliampere per volt for the curve marked glg2=1 l06. The mid-band frequency was L15G-kc. The curves of Figure 5 show the very Wide range of variation of 4characteristic which can be obtained by varying the gain of the Valves, and it will be noted `that as the product gl and y2 is raised, the impedance at the mid-band frequency rises and the curves broaden out until at some value between x10-7 and 9.7 107 the impedance remains practically constant over a range of approximately kc. For values of the product of gI and g2 above 9.7 10-7 the curves exhibit increasingly double-bumped shapes. The curves of Figure 6 show the effects obtained by varying theseries resistance I of Figure 2, these curves being taken for a value of the product of gl and y2 equal to 9.7 10*7 mhos. It will be seen from Figure 6, in which the values of resistance I of Figure 2 are marked on the respective curves, that as this resistance I is increased, the imcreased.

Resistance 2 may also be made variable to conytrol the band Width, the eiectiobtained being that, in general, the bigger this resistance 2 the greater the band width. A practical limit is set at which stray capacity across the resistance 2 produces asymmetry of characteristic. This, however, may be countered by tuning-out such stray capacity with a parallel inductance. Such a parallel inductance is shown in broken lines 1n Figure 2 at L, the stray capacity being also lshown in broken lines.

The same general principles above described as employed to provide improved Wide band pass amplifiers in accordance with this invention, may be employed to provide narrow band highly selective band pass filters with flat topped and steeply sided characteristics. In such an emin the network to be reversed, two (or, if desired, in a more complex -casemore than two) tuned circuits resonant, depending on the circuit employed, either at the mid-band frequency or respectively at two freparallel quencies closely adjacent and symmetrically dis'- posed on either side of the mid-band frequency. There may be included in the platecircuit of the first valve and as shown in Figure 13, a parallel tuned circuit in parallel with a series tuned circuit both circuits being resonant at the midbandr frequency fo. Again the equivalent of the netwerk just described may be employed, this equivalent comprising as shown in Figure 7 two parallel tuned circuits in series with one another and resonant respectively above and below the mid-band frequency at the desired cut-01T frequencies. In the network of Figure '7 one circuit is resonant at one cut-off frequency FI and the other at the other F2, the said frequencies Fl and FZ being closely adjacent and symmetrically disposed on either side of the desired mid-band frequency Fu. The reversed form of the network of Figure '7 is shown in Figure 8. In

Figures 7 and 8 the various elements are identified by the numbers marked thereon, the same convention of a superimposed lN being employed to indicate reversal as in Figures 3 and 4. At about the mid-band frequency (Fu) vthe arrangement acts like a negative parallel resonant circuit as represented in Figure 9 and can be made, therefore, to produce compensating action resulting in a at top to the characteristic. As the frequencies Fi and F2 are approached and passed through, the impedance changes become very rapid-and sharp'cut-ofis are obtained.

In practice either series connected parallel tuned circuits (Figure 7) resonant respectively at 'FI and F2 or a parallel tuned circuit and a series tuned circuit both resonant at'Fo and in with one another (Figure 13) may b e used, lit being a matter of convenience which arrangement "is adopted in any particular case.

It will be appreciated that the parallel tuned circuit and series resonant circuit in parallel with one another as in VFigure 13, and resonant at the same frequency is vthe equivalent of a quartz crystal neutralised with a parallel resonant circuit, i. e. a quartz crystal connected across a parallel tuned circuit both crystal and circuit being resonant at the same frequency F0. Accordingly, where desired, a narrow band pass iilter effect can be obtainedby using a crystal network as just described,V as the network which is to be reversed.

Variable selectivity can be obtained by providing for variation either manually or automatically "of the different elements in the circuit.v As -already stated with reference to Figure 2 the resistance l may be varied but .other elements may be arranged for variation either automatically or manually. YThe various possibilities of variation are indicated diagrammatically in the accompanying AFigures 10, 11, and 12, and, if desired, several of these may be used in any one case.

Figures l() and 12 represent embodiments wherein ordinary circuit elements make up the 'network which is to be reversed, while Figure 1l 'shows an embodiment in which a crystal is included in said network which is to be reversed.

In Figures 10, '11, and 12,v which show arrange- 'ments suitable for use in intermediate frequency amplifierathe arrows'and narrow heads associated with velements indicate manual controls which may be provided; the leads Si represent leads upon which there may be superimposed, for auto- `matic control purposes, 4voltages derived in well known lmanner in .dependence upon incoming (desired) 'carrier strength; while kthe references S2 shown beside variable condensers, indicate that these condensers may, if desired, rbe arranged to be automabtically varied (in known manner by means yof a discriminator circuit) in dependence upon the amount of 'interference present at any time. The letters BW in brackets indicate that the controls in question influence band width, and the letters BD in brackets'indicate that the controls in question displace the band in relation to interference.

Although in the speciiic arrangements hereinbefore described, two-valve circuits are described, the invention is not limited to such circuits for the essential feature thereof, namely the obtaining of the necessary reversal effects, can becarried into practice by means of circuits employing ,y

other numbers of valves, e. g. single valve circuits. v

From the point of view of the lampliiier chain, the apparatus' shown in Figures 1o, 11, .and 12, including the twovalves Vl and V2 are additional to the amplifier chain and serve as coupling .elements in the amplifier chain. This is indicated by the arrows marked IF in these iigures. The function of the valves VI and V?. inthe arrangments of Figures 10, 11, and 12 is to modify the impedance connected inthe grid of valve Vi to have certain desired characteristics. Thus an impedance Z1 becomes effectively 1. l ZI gigs-Zn (I) As will be apparent from what follows,'it is lpossible to produce such an impedance within the amplifier chain itself, that is without the additional Valves VI and V2. Figure 14` of the accompanying drawings illustrates such an amplifier chain wherein the impedance Zn is modified by means of the impedance Ziv and` the grid 3 of valve A.

Defining the ratio Z asZ' (a transfer impedance) it may be shown that ZI=E=9A1ZIZIU+ZN i' l yf l gBZH QASZIV (H) which may be rewritten ZI=QA19BZZ111+Z1V Z-H-QBQABZIV- (IH) The denominator of Expression III is similar to that of Expression I and may be considered as the effective admittance of the coupling between valves A and B, that is the reversed characteristic of Ziv is superimposed on Zn. The resulting characteristic is designed to correct that of ZrZm (neglecting ZIV in comparison with Zin). By making Ziv a series resonant circuit and Zn a parallel resonant rcircuit and by correctly relating ge to gAs and the Qs of the circuits,

gBZlI can be made slightly bigger than -gAaZiv at the resonant frequency and the value of can Abe made to hold constant, or to increase, on moving 'from 'the .resonant frequency ,as may .be

required. The feedback action of Zw on to the third grid of valve A can therefore be regarded as a compensating action, which can be used to flatten out the response curve over a band of frequencies: it can virtually make Zu aperiodic and at the same time raise its impedance, or it can make it of any required low Q to obtain a required correction and of high impedance instead of low impedance.

Figures 15a and 16a illustrate two useful circuits involving the above principles.

In the arrangement of Figure 15a the wide band transformers TI and T2 (which can conveniently be of the back-to-back tapered line type and which respectively incorporate Z1 and Z111) can be adjusted to give flat top response, or double hump response, the compensating arrangement provided by the superimposition of ZIV (reversed) onto Z1 lling in the curve and at the same time allowing high gain in the intercoupling stage to be used as shown in Figure 15b wherein the shape of the response due to the Wide band transformers is shown by the dot and dash line, the shape due to the high gain compensated stage is shown by the dash line, and the overall result by the full line.

In the arrangement of Figure 16a the tunings of the input circuit Zr and output circuit Zin tunings are staggered above and below the intermediate frequency Fo so as to produce a gentle double humped curve. The middle circuit Zn tuned to F is a high gain compensated circuit effectively producing the low Q correction required.

Figure 16h shows the characteristics for the .1.

various parts of Figure 16a, the characteristics for the input and output circuits respectively at frequencies Fl and F2 being shown by the dot and dash lines F1 and F2 respectively, the shape due to the high gain compensated stage is shown by the dash line, and the overall result by the full line.

It has been recognized from general considerations, and from experimental evidence, that positive feedback multi-valve arrangements in which negative feedback is introduced either stage by stage or overall, possess advantages as regards stability, smoothness of adjustment, reproducibility, and so forth, and negative feed back can be applied with advantage to any of the reversed impedance circuits disclosed above. In Fig. 14 for example, negative feedback is obtained by means of the unbypassed cathode resistance R for the tubes A and B.

It may easily be shown that the input impedance YIN of the arrangement illustrated in Figure 1'7 is given by the equation Putting Z1 and Z111 each to zero, i. e. with no cathode back coupling, this becomes the familiar Ynv: gAQfBZII If on the other hand cathode back coupling .be introduced to the extent that the terms gAZi and gBZni are each 1; the expression becomes -gAgBZ11. ZU

YIN

low Q, it will be seen that the variation of YIN over a band of frequencies will be practically dependent upon the Variation of Zn. If Zn'be a series circuit, it can therefore be used to compensate a parallel circuit placed across the input terminals as it will be reversed: the reversal effect Awill however become more and more independent of the constants of the valves as gAZr and gBZIn are made bigger and bigger compared with unity. If Zr, Zn and m are each constituted as parallel resonant circuits and are of equal Q values and if Znis made equal to kZI, Z111 being made equal to Z1 (still assuming gAZI and gBZm 1) then It will be seen therefore that the compensation of a parallel tuned circuit placed across the input terminals can be effected by means of three parallel resonant circuits, and, by the correct choice of values, valve constants can be eliminated.

In applying reversed impedance technique to television carrier frequencies, it becomes important to use the form of circuit best adapted to lump-in stray capacities. The fundamental circuit is, in effect, the full line circuit as shown in Figure 18a giving the correction over the region of frequencies required, whilst the dotted line circuit may be regarded both as a means of producing a cut-off action defining the sides of the resonance curve and of tuning out stray capacity. The circuit of Figure 18a is equivalent to that of Figure 18h acting as Figure 18o about the mid-band Fo.

For Wide bands such as for a television spectrum, either at carrier frequency or at an intermediate frequency, it is required to make the impedance at frequency Fo of the equivalent series circuit as high as possible, and at the same time remove lfrequencies: F1 and F2 as far away as possible from Fo.

It has been found that the form of circuit illustrated in Figure 19, which again is equivalent to the fundamental circuit, produces the best results when stray capacity is the circuit limitation.

In this circuit the stray capacity C is made to tune with a coil L to the frequency F2 (the higher frequency) and a relatively tight capacity coupling couples the parallel circuit so constituted to a circuit of the same constants tuned to the same frequency F2.

To produce the requisite separation between F1 and F2 for a 10 megacycle I. F. for the television side band spread, condenser C1 has to be of the order of twenty times that required to produce critical coupling. It is of course obvious that the bigger the Value of C1 the smaller is the coupling reactance, the tighter the coupling, and the bigger the separation between the equivalent frequencies F1 and F2.

In the application of feed back to high frequency ampliners, such as used in television systems, it is necessary to compensate for phase shift in the amplifier due to transit time of electrons.

If the phase shift within the amplifier is proportional to frequency, then the output voltage will be a faithful copy of the input Voltage but will be delayed or advanced in time by a constant -amount for all frequencies. Now, when it is desired to feed back a portion ofthe output voltage to the input, in order to modify the characteristics of the amplifier, then a difference of time between the two voltages obviously cannot be permitted. Steps must therefore be taken to reduce this time difference to a negligible amount.

Electron transit time effects within a valve give rise to a time delay or vnegative phase shift which may amount te as much as 15'? at 10 megacycles per second or 70 at 45 megacycles per second. In order to counteract this effect it is necessary to advance the phase by an equal amount. The negative phase shift is proportional to frequency, consequently the compensating positive phase shift must also be proportional to frequency.

The circuits and description which vfollow .are the outcome of experimental and analytical work carried out on a 10 mc./sec. two valve impedance inverting arrangement which-enables feedback to be applied at very high frequencies over a wider band width than was hitherto possible.

Two-distinct methods can be used. These will be further explained with reference lto Figure 20 which represents two valves Vl and V2 of an amplier, coupled by means of the impedance Z1. The following discussion is concerned with the phase of i2 with respect to e1. The two methods which can be adopted in'order to advance the phase of i2 with respect to ei aretherefore as follows: v l

(i) The phase of i1 can be advanced with respect to e1 and/or the phase of i2 can be advanced with respect to e2.

(ii) The phase of e2 can be advanced with respect to i1.

Firstly methods will be v considered for *advancing the phase of the anode current ofv a valve with respect to the voltage applied Ain th grid circuit. f

In the diagrammatic arrangement shown in.

Figure 21 let e be the voltage applied in the grid circuit of a valve V, Z an impedance through which the anode current can pass but in such a way that the voltage e is not effected, and eg the resulting voltage between grid and cathodeof the valve.

Thus the anode current is equal to ge (the value it would have if Z were not present) multiplied by the phase advancing term This arrangement therefore produces the right result and provided the angles are not too great the change in amplitude can be neglected.

Figures 22 to25 show practical arrangements for producing this result. Figure 22 `is a simple back-coupling with a negative mutual, represented by the reference -M. Figure 23 is similar to-Figure 22 but with a split winding anda neutralising condenser Cn to prevent direct coupling between the anode load and the grid circuit via the stray capacity Cs. Figure 2ii1lustrates a voltage feedback arrangement wherein feedback is effected through the small condenser Cf. This type of feedback 'is dependent upon the impedance of the .anode load, but can be applied very successfully incertain cases. The arrangement -of Figure 25 is similar to that of Figure 22 except that the screen ycurrent is fed back instead of the anode current.

The required result is obtained if a negative inductance be included inthe rcathode lead. Similarly for narrow band widths a simple condenser (shunted by means of `a suitable resistance or inductance to carry the D. C. component) in series with the cathode lead produces a `,positive phase shift in the anode current, with respect to the applied grid voltage.

Figure 26 shows one Way of obtainingjthe leffeet of ra negative inductance in the cathode lead. 'Considering now the second method yand referring to Figure 20 this second method consists in advancing the phase of ez with respectto the previous anode current i1. 'l

-In many cases the coupling impedance Z1 of Figure 20 would be Ya damped parallel circuit. If this circuit be .arranged as shown in Figure 27 then it can be written: y

,e :i (T4-Z2) Za 2 lf-i-Zc-i'za But l Y 1-|-Z2+Z3=r(1+7'K) Where f ZQAf y I? f and vZ3';j.dLr Therefore the large values of damping resistance used in television circuit, the `term (l-l-jeC) can beV made to produce ample phase shift.

By using the circuit of Figure 28 for Z1 of Figure 20 the phase shift and damping can be controlled independently by controlling the ratio of r1 to r.

Other types of complexucircuits may be employed. Thus a circuit, which possesses special properties as regards maintaining an input impedance which fluctuates very slowly over a relatively Wide band of frequencies, and at the same time has stray capacities which in the ordinary way vitiate the results, is illustrated in Figure 29. The sign and the value of the mutual inductance M is such that the equivalent circuit is indicated in Figure 30 where the series circuits SI, SII and SIII and the parallel tuned circuits PI, PII and PIII are each tuned to the same frequency and are so proportioned as to obtain the desired impedance conditions.

The circuits of Figures 31 and 32 possess certain conveniences of adjustment and enable the valves which produce the compensation effect to be part of the amplifying chain. Figure 31 produces variable selectivity as the feed back potentiometer P is altered, whilst Figure 32 may be adjusted to obtain high gain with wide band width.

It should be understood that the grids labelled I and 3 in both figures can be interchanged, whilst the grid of valve VI in Fig. 32 may, in-

stead of being connected as shown, be connected to the point A.

In lFigure 32 the input impedance to bandcoupled circuit III is of inverse nature to that of circuit II thus producing a compensation action which in conjunction with the other circuit actions produces the desired result.

What we claim is:

1. An electric circuit comprising first and second electron discharge tubes, an impedance included in the grid circuit of the first tube, and

a second impedance coupling the plate of the rst tube to the grid of the second, characterized in that the circuit constants, the impedances and the operating parameters of the tube are so chosen that the coupling impedance between the tubes is reversed and superimposed on and substantially neutralizes the grid circuit impedance of the first tube.

2. A wide band amplifier circuit comprising first and second electron discharge tubes, an impedance included in the grid circuit ofthe first tube, a second impedance coupling the plate of the first tube to the grid of the second and a capacity coupling the anode of the second tube to the grid of the first, characterised in that the circuit constants, the impedances and the operating parameters of the tubes are so chosen that the coupling impedance between the tubes is reversed and superimposed on and neutralises the grid circuit impedance of the first tube.

3. A wide band amplier as defined in claim 2 wherein the impedance included in the grid circuit of the first tube is constituted by a parallel tuned circuit and the coupling between the tubes consists of a first resistance in'series with a second resistance which latter resistance is shunted by a series circuit comprising a third resistance, a capacity, and an inductance.

4. A wide band amplifier as defined in claim 2 wherein the second coupling impedance consists of a variable resistance in series with a parallel tuned4 circuit and a piezoelectric crystal in shunt to the tuned circuit.

5. A wide band amplifier, circuit as defined in claim 2 wherein means are provided for obtaining negative feed-back.

6. A wide band amplifier circuit as defined in claim 2 wherein the capacity coupling is provided for obtaining feed-back from the output to the input of the amplifier, and wherein means are included in the second impedance for correcting for phase shift due to electron transit time whereby the energy fed back is substantially brought into phase with the input.

'7. A band-pass amplifier circuit arrangement comprising first and second electron discharge tubes, the first including at least a control grid and an additional grid, an impedance included in the controlr grid circuit of the first tube, a second impedance coupling the plate of the first tube to the grid of the second, two series connected impedances included in the plate circuit of the second tube, and a connection from the junction pointiof the two series connected impedances to said additional grid in the first tube, the arrangement being such that the reversed characteristic of the series connected impedance remote from the anode of the second tube is superimposed on said secondA impedance to produce a resultant characteristic which corrects the characteristic of the remaining one of the series connected impedances and the impedance in the grid circuit of the first tube.

8. A wide band amplifier circuit as dened in claim 7 wherein means are provided for obtaining negative feed-back.

9. A wide band amplifier circuit as defined in claim '7 wherein means are provided for obtaining feed-back from the output to the input of the amplifier, and wherein means are included in the second impedance for correcting for phase shift due to electron transit time whereby the energy fed back is brought into phase with the input.

10. A vacuum tube circuit comprising first and second tubes, a first network connected to the input of the first tube, a second network coupling the output of the rst tube to the input of the second, and means for feeding energy from the output of the second tube to said first network to thereby parallel the self admittance of the first network with an effective impedance proportional to the admittance of the second network but reversed in sign, whereby the admittance characteristic of the firstnetwork is altered in accordance with the negative of the admittance of the second network.

1l. A vacuum tube circuit as defined in claim 10 including means for advancing the phase of currents traversing said tubes to compensate `for the time lag occurring in said tubes.

12. A vacuum tube circuit as defined in claim 10 including negative feed back means associated.

with at least one tube to render the action thereof less dependent on variation of the transconducty, ance thereof.

to thereby parallel the self admittance of the rst network with an effective impedance proportional to the admittance of the second network but reversed in sign, whereby the admittance characteristic of the first network is modified in accordance with negative of the admittance of the second network.

NOEL MEYER RUST. JOSEPH DOUGLAS BRAILsFoRD. ERNEST FREDERICK GOODENOUGH. 

